Chopped Hall effect sensor

ABSTRACT

A chopped Hall effect sensor topology includes a switched Hall plate, an amplifier responsive to an output of the switched Hall plate and a filter stage responsive to the output of the amplifier and including an anti-aliasing filter and a selective filter that is tuned to the modulation frequency. The switched Hall plate includes a Hall element and a Hall plate modulation circuit that modulates the Hall offset signal component or the magnetic signal component. In embodiments in which the Hall offset signal component is modulated by the switched Hall plate, the amplifier, if chopped, includes an even number of additional modulation circuits. In embodiments in which the magnetic signal component is modulated by the switched Hall plate, the amplifier contains an odd number of modulation circuits. The described topology provides a low noise, fast response time Hall effect sensor.

CROSS REFERENCE TO RELATED APPLICATIONS

Not Applicable.

STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH

Not Applicable.

FIELD OF THE INVENTION

This invention relates generally to Hall effect sensors and, moreparticularly, to a chopped Hall effect sensor having fast response timeand reduced noise.

BACKGROUND OF THE INVENTION

Hall effect sensors are used in a wide variety of applications includingindustrial and consumer applications. As one example, Hall effectsensors are widely used in the automotive industry for mechanicalposition sensing, such as gear tooth sensors used in brake systems. Suchapplications require accuracy.

Hall effect elements or plates experience imbalances due to resistancegradients, geometrical asymmetries and piezoresistive effects which canintroduce an offset voltage. The magnitude and polarity of the offsetvoltage are a function of stresses in the semiconductor from which theelement is formed, which stresses vary with mechanical pressure andtemperature. Various techniques have been used to address and cancel theHall offset voltage, including chopper stabilization techniques.

One type of chopped Hall effect sensor includes a switched Hall plate, achopped amplifier, and a low pass filter. The switched Hall plate,sometimes referred to alternatively as a spinning Hall plate, includes aHall element having (typically) four contacts and a modulation switchcircuit to periodically connect the supply voltage and the amplifierinput to one pair of contacts or the other. Quadrature phases ofoperation are defined by complementary clock signals. Use of such aswitched Hall plate provides a way to discriminate the Hall offsetvoltage (referred to herein as the Hall offset signal component) fromthe magnetically induced signal (referred to herein as the magneticsignal component). In one such circuit, the switched Hall platemodulates the magnetic signal component and the offset signal componentremains substantially invariant. The chopped amplifier demodulates themagnetic signal component and modulates the offset signal componentwhich is then attenuated by the low pass filter to provide the sensoroutput signal. While this technique is effective to remove the Halloffset voltage, the resulting ripple on the sensor output signal and thesensor response time must be balanced since, the more filtering applied,the lower the resulting ripple, but also the slower the sensor responsetime.

Some more recent Hall effect sensor applications additionally requirefaster response times to input magnetic field steps. As one example,Hall effect sensors used in current sensing applications must respondquickly to step changes in the magnetic field, for example in order torapidly detect fault conditions, such as short circuits in automobilebatteries.

One chopped Hall effect sensor that improves upon the above-describedsensor in terms of response time is described in U.S. Pat. No. 5,621,319entitled “Chopped Hall Sensor with Synchronously Chopped Sample and HoldCircuit” which issued on Apr. 15, 1997 to Allegro Microsystems, Inc. ofWorcester, Mass., the Assignee of the subject invention. The describedsensor includes a switched Hall plate and an amplifier, with theswitched Hall plate arranged to modulate the magnetic signal componentand maintain the offset signal component substantially invariant. Here,the modulated magnetic signal component is demodulated by sample andhold techniques. According to this technique, signal demodulation isperformed by tracking and holding during both clock phases and theninverting the modulated signal during the second phase. In this way,this circuit entirely eliminates ripple on the sensor output signal andthus, provides a faster step response time by avoiding ripple filtering;however, these benefits are achieved at the cost of a degraded signal tonoise ratio. This is because the sampling and holding operation canproduce noise fold back (i.e., aliasing) since the baseband noise isundersampled.

SUMMARY OF THE INVENTION

A Hall effect sensor according to the invention includes a Hall element,a Hall plate modulation circuit, an amplifier, and a filter including aselective filter tuned to the modulation frequency. The Hall platemodulation circuit is responsive to the output signal of the Hallelement and operates to modulate the magnetic signal component or theoffset signal component of the Hall output signal. The amplifier isresponsive to the modulation circuit output signal and provides anamplifier output signal to the filter. The filter includes ananti-aliasing filter coupled between the amplifier and the selectivefilter.

With this arrangement, the Hall effect sensor is provided with fastresponse time through the use of the selective filter that removes theoffset signal component with its associated ripple, thereby eliminatingthe significant low pass filtering requirements of some conventionalchopped Hall effect sensors. A high signal to noise ratio is achievedwith the use of the anti-aliasing filter that removes frequencycomponents above a predetermined frequency, so that the selective filtermeets the Nyquist criterion for noise signals, thereby reducing basebandnoise by preventing aliasing.

In embodiments in which the offset signal component is modulated by themodulation circuit, an even number of additional modulation circuits areprovided between the output of the Hall plate modulation circuit and theinput to the filter. In this way, the offset signal component ismodulated when it reaches the selective filter for removal.Alternatively, in embodiments in which the magnetic signal component ismodulated, the amplifier includes an odd number of additional modulationcircuits between the output of the Hall plate modulation circuit and theinput to the filter, again ensuring that the offset signal component ismodulated when it reaches the selective filter for removal.

Embodiments of the invention may include one or more of the followingfeatures. The amplifier may be a closed or open loop amplifier. Inclosed loop embodiments, the loop may be closed at the input to theanti-aliasing filter, at the output of the anti-aliasing filter, or atthe output of the selective filter. Also, in embodiments in which theamplifier loop is closed at the output of the anti-aliasing filter, theanti-aliasing filter may serve an additional loop compensation purpose.The filter may include a smoothing filter.

In embodiments in which the offset signal component is modulated by themodulation circuit, the amplifier may or may not be chopped. However, inembodiments in which the magnetic signal component is modulated by themodulation circuit the amplifier must be chopped. In embodiments inwhich the amplifier is chopped and the amplifier is a closed loopamplifier, the feedback network may or may not be chopped.

One illustrative selective filter includes a plurality of sample andhold circuits arranged in pairs, with each sample and hold circuithaving an input coupled to the output of the anti-aliasing filter and anoutput. The filter further includes an averaging circuit having aplurality of inputs, each coupled to the output of a respective sampleand hold circuit, and an output at which the selective filter outputsignal is provided. Each of the sample and hold circuits samples theinput signal at the modulation frequency and at a phase separated fromthe phase of the other sample and hold circuit of the same pair by 180degrees and at a phase arbitrarily separated from the phase at which theother pairs of sample and hold circuits operate.

In one particular embodiment, the anti-aliasing filter includes a firstsample and hold circuit having an input coupled to the output of theanti-aliasing filter, a second sample and hold circuit having an inputcoupled to the output of the anti-aliasing filter, and an averagingcircuit having inputs coupled to the outputs of the first and secondsample and hold circuits. The first sample and hold circuit samples theinput signal at times t=t0+N·TSF and the second sample and hold circuitsamples the input signal at times t=t0+(N+1)·TSF, wherein t0 is anarbitrary time, N is an integer and TSF is 1/(2·fCLK), where f_(CLK) isthe modulation frequency.

BRIEF DESCRIPTION OF THE DRAWINGS

The foregoing features of the invention, as well as the invention itselfmay be more fully understood from the following detailed description ofthe drawings, in which:

FIG. 1 is a block diagram of a chopped Hall effect sensor according tothe invention;

FIG. 2 shows a conventional switched Hall plate for use in the sensor ofFIG. 1 to modulate the Hall offset signal component;

FIG. 2A shows clock signals for switched Hall plate of FIG. 2;

FIG. 2B shows an illustrative modulated Hall offset signal componentprovided by the switched Hall plate of FIG. 2;

FIG. 2C shows an illustrative magnetic signal component provided by theswitched Hall plate of FIG. 2;

FIG. 3 shows a conventional switched Hall plate for use in the sensor ofFIG. 1 to modulate the magnetic signal component;

FIG. 3A shows clock signals for the switched Hall plate of FIG. 3;

FIG. 3B shows an illustrative Hall offset signal component provided bythe switched Hall plate of FIG. 3;

FIG. 3C shows an illustrative modulated magnetic signal componentprovided by the switched Hall plate of FIG. 3;

FIG. 4 is a block diagram of one embodiment of the chopped Hall effectsensor of the invention in which the switched Hall plate modulates theHall offset signal component;

FIG. 4A shows an illustrative Hall output signal with the Hall offsetsignal component modulated as provided at the output of the switchedHall plate of FIG. 4 and also shows the magnetic signal component of theHall output signal;

FIG. 4B shows the signal provided at the input to the gain stage of FIG.4 having the offset signal component demodulated and the magnetic signalcomponent modulated and also shows the demodulated offset signalcomponent;

FIG. 4C shows the signal provided at the input to the filter stage ofFIG. 4 having the offset signal component modulated and the magneticsignal component demodulated and also shows the demodulated magneticsignal component;

FIG. 4D shows the filtered signal provided at the output of theanti-aliasing filter of FIG. 4 and the demodulated magnetic signalcomponent;

FIG. 4E shows the signal provided at the output of the selective filterof FIG. 4;

FIG. 5 is a block diagram of an alternative chopped Hall effect sensoraccording to the invention in which the switched Hall plate modulatesthe magnetic signal component;

FIG. 5A shows an illustrative Hall output signal with the magneticsignal component modulated as provided at the output of the switchedHall plate of FIG. 5 and also shows the offset signal component of theHall output signal;

FIG. 5B shows the signal provided at the input to the filter stage ofFIG. 5 having the offset signal component modulated and the magneticsignal component demodulated and also shows the demodulated magneticsignal component;

FIG. 5C shows the signal provided at the output of the anti-aliasingfilter of FIG. 5 and the demodulated magnetic signal component;

FIG. 5D shows the signal provided at the output of the selective filterof FIG. 5;

FIG. 6 shows an illustrative feedback network embodiment for theamplifiers of FIGS. 1, 4, and 5;

FIG. 7 shows an illustrative embodiment for the selective filter ofFIGS. 1, 4, and 5; and

FIG. 7A shows the transfer function of the selective filter of FIG. 7;

FIG. 7B shows the signals of FIG. 4D with notations to illustrateoperation of the selective filter of FIG. 7;

FIG. 8 is a block diagram of a farther alternative chopped Hall effectsensor according to the invention in which the switched Hall platemodulates the Hall offset signal component;

FIG. 9 shows an illustrative waveform representing a magnetic stepdisturbance;

FIG. 9A shows illustrative output signals of the low pass filter of theinventive chopped Hall effect sensor in response to the step disturbanceof FIG. 9, one where a Hall offset voltage exists and one with no Halloffset voltage; and

FIG. 9B shows the response of the selective filter of the inventivechopped Hall effect sensor to the input step disturbance of FIG. 9.

DETAILED DESCRIPTION OF THE INVENTION

Referring to FIG. 1, a chopped Hall effect sensor 10 includes a switchedHall plate 14 providing a switched Hall output signal 16, an amplifierstage 24 having an input responsive to the switched Hall output signal16 and an output at which is provided an amplifier output signal 26, anda filter stage 34 having an input responsive to the amplifier outputsignal 26 and an output at which is provided a sensor output signal 36.The filter stage 34 includes an anti-aliasing filter 38 and a selectivefilter 40, as shown.

With this arrangement, the Hall effect sensor 10 is provided with lownoise and fast response time. Fast response time is achieved with theuse of the selective filter 40 that removes the offset signal componentwith its associated ripple, thereby eliminating the significant low passfiltering requirements of some conventional chopped Hall effect sensors.A high signal to noise ratio is achieved with the use of theanti-aliasing filter 38 to remove frequency components above apredetermined frequency, so that the selective filter 40 meets theNyquist criterion for noise signals, thereby reducing resulting basebandnoise by preventing aliasing. In the illustrative embodiment, theanti-aliasing filter 38 removes frequency components above the sensorclock frequency and the selective filter 40 samples at twice the clockfrequency. In a preferred embodiment, the anti-aliasing filter is a lowpass filter. Notably the filter requirements of the anti-aliasing filterare relaxed as compared to conventional chopped Hall sensors in whichthe low pass filter is the mechanism for removing the offset signalcomponent with its associated ripple. In one illustrative embodiment,the anti-aliasing filter 38 is a first order low pass filter.

The switched Hall plate 14 includes a Hall element or plate 18 having anoutput at which is provided a Hall output signal 20 that varies inaccordance with a sensed magnetic field and a Hall plate modulationswitch circuit, or simply a Hall plate modulation circuit 22 having aninput responsive to the Hall output signal and an output at which isprovided the switched Hall output signal (also referred to herein as themodulation circuit output signal) 16. The switched Hall output signal 16is coupled to an input of the amplifier stage 24, as shown. The Halloutput signal 20 and switched Hall output signal 16 include a magneticsignal component V_(H) and a Hall offset signal component V_(op).

As will be described, the modulation circuit 22 may be controlled tomodulate the Hall offset signal component V_(op) or the magnetic signalcomponent V_(H) at a modulation frequency, referred to alternativelyherein as the clock frequency f_(CLK). An illustrative, conventionalmodulation circuit that modulates the Hall offset signal componentV_(op) is shown and described in connection with FIGS. 2-2C and anillustrative, conventional modulation circuit that modulates themagnetic signal component V_(H) is shown and described in connectionwith FIGS. 3-3C.

In embodiments in which the Hall plate modulation circuit 22 modulatesthe Hall offset signal component V_(op) the Hall effect sensor 10includes an even number of modulation circuits between the output of theHall plate modulation circuit 22 and the input of the filter stage 34 inorder to enable the selective filter 40 to remove the modulated offsetsignal component. One illustrative Hall effect sensor of this type isshown in FIG. 4. With this arrangement, the even number of modulationcircuits in the amplifier stage operate to, one or more times,demodulate and again modulate the offset signal component to ensure thatthe offset signal component is modulated when it reaches the filterstage and thus, can be removed by the selective filter 40 in order torecover the magnetic signal component. In fact in such embodiments, theamplifier stage 24 may not be a chopped amplifier and thus, may notperform any signal modulation, thereby resulting in zero modulationcircuits between the output of the Hall plate modulation circuit 22 andthe input to the filter stage 34.

In embodiments in which the Hall plate modulation circuit 22 modulatesthe magnetic signal component V_(H), the Hall effect sensor 10 includesan odd number of modulation circuits between the output of the Hallplate modulation circuit 22 and the input of the filter stage 34 inorder to enable the selective filter 40 to remove the offset signalcomponent. One illustrative Hall effect sensor of this type is shown inFIG. 5. With this arrangement, the odd number of modulation circuits inthe amplifier stage ensures that the offset signal component ismodulated when it reaches the filter stage and thus, can be removed bythe selective filter 40 in order to recover the magnetic signalcomponent. As will become apparent, at least one amplifier 30 a-30 ncomprising the amplifier stage 24 is a chopped amplifier in embodimentsin which the modulation circuit 22 modulates the magnetic signalcomponent since otherwise, the offset signal component would not beup-converted for removal by the selective filter (and optionally morethan one amplifier 30 a-30 n is chopped, as long as an odd number ofchopper stages are used).

The amplifier stage 24 may include one or more amplifiers 30 a-30 n.Each amplifier 30 a-30 n has a gain stage 32 a-32 n and may or may notbe chopped. If chopped, the amplifier includes at least one modulationcircuit, as may be provided by a pair of cross-coupled switches 46 a-46n and may additionally include a second modulation circuit, as may beprovided by a pair of cross-coupled switches 42 a-42 n, as shown indotted lines for amplifier 30 a and in solid lines for amplifier 30 n.Each of the amplifiers 30 a-30 n, like the Hall element 18, has anassociated offset voltage, here shown by respective voltage sources 48a-48 n at the input to the respective gain stage 32 a-32 n. Themodulation circuits 42 a-42 n and 46 a-46 n operate to modulate ordemodulate the processed signal and may be implemented, for example withMOSFET switches. In the illustrated embodiment, the modulation circuits42 a-42 n and 46 a-46 n operate at the clock frequency f_(CLK). It willbe appreciated however that the amplifier stage 24 may be chopped at adifferent frequency than the clock frequency used by the Hall platemodulation circuit 22.

The particular choice of number of amplifiers 30 a-30 n comprising theamplifier stage 24 is based generally on the desired overall gain forthe amplifier 24. It is desirable that the amplifier gain be largeenough so that any offset associated with any non-chopped circuitry thatfollows the filter stage 34 is significantly less than the Hall andamplifier offsets, thereby minimizing the contribution of such“back-end” offset. Once an overall gain is selected, how much gain isprovided by any given amplifier 30 a-30 n requires consideration ofvarious factors such as bandwidth and response time. For example, inclosed loop configurations, the higher the gain for a particular stage,the lower the bandwidth; however, the lower the bandwidth, the slowerthe sensor response time. It will be appreciated that in embodimentscontaining more than one amplifier 30 a-30 n, each amplifier need not beidentical in terms of topology and specifications. For example, the gainof the different amplifiers 30 a-30 n may vary. Also the topologies mayvary. For example, the amplifiers 30 a-30 n may be closed or open loopamplifiers.

Whether one or more of the amplifiers 30 a-30 n includes one or twomodulation circuits 42 a-42 n, 46 a-46 n is based on whether the Hallplate modulation circuit 22 modulates the Hall offset signal componentV_(op) or the magnetic signal component V_(H), since as mentioned above,in the former case, an even number of modulation circuits is requiredbetween the output of the Hall plate modulation circuit 22 and the inputof the filter stage 34 in order to cancel the offset signal componentand in the latter case, an odd number of modulation circuits is requiredbetween the output of the Hall plate modulation circuit and the input ofthe filter stage in order to cancel the offset signal component.

As mentioned above, the anti-aliasing filter 38 removes frequencycomponents as necessary to ensure that the selective filter 40 meets theNyquist criterion for noise signals. In the illustrative embodiment, inwhich the selective filter 40 samples at a frequency of 2 f_(CLK), thefilter 38 has a cutoff frequency on the order of 0.35 f_(CLK). As willbe appreciated by those of ordinary skill in the art, various filterdesigns are possible for the low pass filter and the particular cutofffrequency of the filter 38 is a function of the sampling frequency ofthe selective filter 40 and the desired sensor response time.

The selective filter 40 is tuned to a frequency at which incomingsignals are eliminated and also attenuates other frequency componentsabove a given bandwidth. In particular, the selective filter 40 is tunedto a frequency selected to ensure removal of the offset signal componentwith its associated ripple. Thus, the selective filter is tuned to theclock frequency f_(CLK) at which the offset signal component ismodulated when it reaches the selective filter.

In the illustrative embodiment, the selective filter 40 is a sampleddata filter in the form of a sinc filter having a frequency domaintransfer function shaped like a sinc function (see FIG. 7A) and a timedomain transfer function shaped like a rectangular function. Theselective filter 40 is a discrete time filter in the sense that itszeros in the frequency domain are at exactly the harmonics related toone-half of the sample frequency. In one preferred embodiment, theselective filter samples at a frequency equal to twice the sensor clockfrequency f_(CLK). Thus, in this embodiment, the zeros are located atN(f_(SF)/2) or Nf_(CLK), where N is any integer. Thus, the filter 40removes all the signal components at the clock frequency f_(CLK) and itsharmonics and attenuates the other frequency components above a givenbandwidth. In this way, the selective filter eliminates the undesiredoutput ripple, whose amplitude is proportional to the DC input offsetsignal component. Thus, the resulting sensor output signal 36 includesonly the magnetic signal component, since the selective filter entirelyeliminates the offset signal component with its associated ripple. Theextent of attenuation at frequencies other than harmonics of the clockfrequency is a function of the number of samples of the input signaltaken within one clock period, with greater attenuation achieved bytaking more samples. It will be appreciated by those of ordinary skillin the art that the particular selection of the sample frequencyrequires a tradeoff between greater signal attenuation as is achieved byaveraging a larger number of samples (as will provide a higher overallsampling frequency) versus area efficiency achieved by using the minimumnumber of samples (as will provide a lower overall sampling frequency).

It will be appreciated by those of ordinary skill in the art that whilethe illustrative selective filter 40 is a sample based sinc filter,alternative filter designs are possible while still achieving thebenefits of the illustrative selective filter; namely, of eliminatingthe offset signal component with its associated ripple. In a preferredembodiment, the selective filter 40 is an averaging filter and may takethe form of a continuous time filter, discrete time filter, an analogfilter, or a digital filter. As one example, a continuous time combfilter may be used.

The filter stage 34 may include an optional smoothing filter 44 tofurther attenuate high frequency content (not located at the clockharmonics) in addition to the attenuation already supplied by theselective filter 40. Additionally, since this smoothing filter 44 isonly intended for high frequency attenuation (secondary side lobes ofthe selective filter transfer function), its cutoff frequency does notneed to be small. For example, in the illustrative embodiment, thesmoothing filter has a cutoff frequency of at least f_(CLK). Thus, thesmoothing filter 44 does not introduce any significant delay to thesensor.

Referring also to FIG. 2, a conventional switched Hall plate 50 of thetype that modulates the Hall offset signal component is shown to includea Hall element or plate 52 and a Hall plate modulation switch circuit54. The Hall element 52 includes four equally spaced contacts 52 a, 52b, 52 c, and 52 d, each coupled to a first terminal of a respectiveswitch 56 a, 56 b, 56 c, and 56 d, as shown. A second terminal ofswitches 56 b and 56 c are coupled to provide the positive node of theswitched Hall output signal 16, here labeled Vo+, and the secondterminal of switches 56 a and 56 d are coupled to provide the negativenode of the switched Hall output signal 16, here labeled Vo−.

Additional switches 60 a, 60 b, 60 c, and 60 d are arranged toselectively couple the Hall contacts 52 a, 52 b, 52 c, 52 d to thesupply voltage Vs and ground. More particularly, switches 56 b, 56 d, 60a, and 60 c are controlled by a clock signal CLK and switches 56 a, 56c, 60 b, and 60 d are controlled by a complementary clock signal CLK/,as shown. The clock signals CLK and CLK/ have two states, a Φ_(0°) stateand a Φ_(90°) state, as shown in FIG. 2A.

In operation, during phase Φ_(0°), current flows from terminal 52 a to52 c and the switched Hall output signal Vo is equal to V_(H)+V_(op),where V_(op) is the Hall plate offset voltage or Hall offset signalcomponent and V_(H) is the magnetic signal component. During phaseΦ_(90°), current flows from terminal 52 b to 52 d and the switched Halloutput signal Vo is equal to V_(H)-V_(op). Thus, the modulation switchcircuit 54 modulates the Hall offset signal component V_(op), as shownin FIG. 2B for zero Gauss. The magnetic signal component V_(H) remainssubstantially invariant, as shown in FIG. 2C.

Referring also to FIG. 3, an alternative conventional switched Hallplate 70 of the type that modulates the magnetic signal component isshown to include a Hall element 72 and a Hall plate modulation switchcircuit 74. The Hall element 72 is identical to element 52 of FIG. 2 andincludes four contacts 72 a, 72 b, 72 c, and 72 d, each coupled to afirst terminal of a respective switch 76 a, 76 b, 76 c, and 76 d. Asecond terminal of switches 76 a and 76 b are coupled to provide thepositive node of the switched Hall output signal, here labeled Vo+, andthe second terminal of switches 56 c and 56 d are coupled to provide thenegative node of the switched Hall output signal, here labeled Vo−.Thus, a comparison of FIGS. 2 and 3 reveals that the output contacts ofthe Hall element are interchanged during the Φ_(90°) phase.

Additional switches 80 a, 80 b, 80 c, and 80 d are arranged toselectively couple the Hall contacts 72 a, 72 b, 72 c, and 72 d to thesupply voltage Vs and ground. Switches 76 b, 76 d, 80 a, and 80 c arecontrolled by clock signal CLK and switches 76 a, 76 c, 80 b, and 80 dare controlled by complementary clock signal CLK/, as shown. Clocksignals CLK and CLK/ are identical to like signals in FIG. 2 and thushave two states Φ_(0°) and Φ_(90°), as shown.

In operation, during phase Φ_(0°), current flows from terminal 72 a to72 c and the switched Hall output signal Vo is equal to V_(H)+V_(op).During phase Φ_(90°), current flows from terminal 72 b to 72 d and theswitched Hall output signal Vo is equal to −V_(H)+V_(op). Thus, themodulation switch circuit 54 modulates the magnetic signal component toprovide a modulated magnetic signal component V_(H), as shown in FIG. 3Cfor zero Gauss. The Hall offset signal component V_(op) remainssubstantially invariant as is shown in FIG. 3B.

It is noteworthy that in the switched Hall plate 50 of FIG. 2, the Halloffset voltage can be represented as a voltage source 58 between theHall element 52 and the Hall plate modulation switch circuit 54. Thus,the Hall offset voltage 58 is added to the amplifier offset voltage 48 afor example (FIG. 1) after being modulated by the Hall plate modulationcircuit 54 (FIG. 2) and demodulated by the modulation circuit 42 a (ifthe same clock frequency f_(CLK) is used for the Hall element and theamplifier). In contrast, in the switched Hall plate 70 of FIG. 3, theHall offset voltage appears at the output of the switched Hall plate, asshown by voltage source 78. Thus, the Hall offset voltage 78 isindistinguishable from the amplifier offset voltage 48a for example inembodiments not including optional modulation circuit 42 a (FIG. 1).Thus, in both cases, the Hall offset voltage and the amplifier offsetvoltage will be simultaneously processed and cancelled by the sensor 10of the present invention.

Referring to FIG. 4, a chopped Hall effect sensor 100 according to theinvention includes a switched Hall plate 50 of the type shown in FIG. 2that provides a switched Hall output signal 114 comprising a modulatedHall offset signal component V_(op) and a substantially invariantmagnetic signal component V_(H), as shown in FIG. 4A. Also shown indotted lines in FIG. 4A is the substantially invariant magnetic signalcomponent of the signal 114. The sensor 100 further includes a choppedamplifier 110 having an input coupled to the output of the switched Hallplate 50 and an output at which an amplified signal 116 is provided. Afilter stage 120, like filter stage 34 of FIG. 1, has an input coupledto the output of the amplifier stage 110 and an output at which thesensor output signal 118 is provided. The filter stage 120 is shownwithout the optional smoothing filter (labeled 44 in FIG. 1).

The amplifier 110 is a closed loop amplifier having a feedback network124, as shown. One illustrative embodiment of the feedback network 124is shown and described in connection with FIG. 6. Use of a closed loopamplifier is desirable due to the resulting high linearity and gainstability over frequency, temperature, process and power supply levels.Furthermore, because the magnetic signal component V_(H) (FIG. 4A) is atbaseband, the tighter bandwidth required of the closed loop amplifier110 does not adversely impact recovery of the magnetic signal component.An additional advantage to the closed loop amplifier 110 in theembodiment of FIG. 4 is the ability to provide a higher gain amplifierwhile maintaining the same gain bandwidth product, therefore reducingthe closed loop bandwidth as much as might be necessary to achievestability.

Depending on the type of feedback network used, significant area savingsmay be achieved in the filter stage 34. For example, if a Millercompensation scheme is used, where the “reflected” capacitance sets thecutoff frequency, and the bandwidth is able to be set to achieve bothloop stability and to allow the selective filter 150 to meet the Nyquistcriteria, then the low pass filter 144 can perform the functionality ofboth the Miller stage and the anti-aliasing filter. Alternatively, ifthe bandwidth cannot be set to meet both the stability and anti-aliasingrequirements, regardless of whether the compensation scheme includes aMiller stage, then a separate anti-aliasing filter must be provided andthe feedback loop can be closed before the anti-aliasing filter.

The switched Hall output signal 114 is coupled to an input of a summingnode 126 and the feedback network 124 is also coupled to an input of thesumming node 126, as shown. The summing node 126, like others describedherein, may be a current or a voltage summing node. The output of thesumming node 126 is coupled to a first modulation circuit, here shown inthe form of a pair of cross-coupled switches 130 that modulate theincoming signal at the clock frequency f_(CLK). The output signal 132 ofmodulation circuit 130 thus contains a modulated magnetic signalcomponent and a demodulated offset signal component, as shown in FIG.4B, Also shown in dotted lines in FIG. 4B is the demodulated offsetsignal component 132 a. Note that the offset signal component of thesignal 132 includes the Hall offset signal component V_(op) (asrepresented by voltage source 58 in FIG. 2) and the amplifier offsetsignal component V_(oa) (as represented by voltage source 134 in FIG.4).

In order to obtain a fast sensor step response time, the clock frequencyf_(CLK) is selected such that the clock period is in the order ofone-half (or less) of the desired step response time (SRT). In oneillustrative embodiment in which the desired step response time is onthe order of 2.0 μs, the clock frequency is on the order of 1 MHz.

Gain stage 138 provides an amplified signal to a further modulationcircuit, here shown in the form of a pair of cross-coupled switches 140,as shown. The gain stage 138 must have a bandwidth large enough to passthe modulated magnetic signal component. In one illustrative embodiment,the gain stage bandwidth is at least five times the clock frequencyf_(CLK). Thus, the tighter bandwidth required to implement the amplifier110 in a closed loop form must be balanced with the minimum bandwidthnecessary to pass the desired magnetic signal component. Since themodulated magnetic signal component does not go through the filter 144or the Miller feedback stage 124, the closed loop bandwidth does notaffect the modulated signal bandwidth. Only the amplifier's sectionthrough which the modulated magnetic signal component passes needs tohave enough bandwidth to pass the desired magnetic signal component.

Modulation circuit 140 operates at the clock frequency f_(CLK) toprovide the amplified signal 116 containing a demodulated magneticsignal component and a modulated offset signal component, as shown inFIG. 4C. Also shown in dotted lines in FIG. 4C is the demodulatedmagnetic signal component 116 a. Note again that the offset beingmodulated by the modulation circuit 140 includes the Hall offset and theamplifier offset which are added at the output of the first modulationcircuit 130.

The amplified signal 116 is coupled to the filter stage 120 and moreparticularly, to the anti-aliasing, low pass filter 144, as shown.Recall that the purpose of the filter 144 is to perform an anti-aliasingfunction by removing frequency components that would fold-back to thebaseband. In the illustrative embodiment, the selective filter 150samples at a frequency equal to twice the clock frequency f_(CLK). Thus,in order to perform its anti-aliasing function, the cutoff frequency ofthe filter 144 must be limited to a maximum of the clock frequencyf_(CLK) and in one illustrative embodiment is on the order of 0.35f_(CLK).

The low pass filter 144 provides the filtered signal 148 of FIG. 4D thatincludes a partially attenuated modulated offset signal component and ademodulated magnetic signal component. Thus, the signal 148 containssuccessive alternating polarity exponential responses which is thechopped amplifier's residual ripple. The degree of attenuation of theripple depends directly on the cutoff frequency of the filter 144. Alsoshown in dotted lines in FIG. 4D is the demodulated magnetic signalcomponent 148 a.

In order to obtain a fast response time, the time constant τ of thefilter 144 must be such that the rise time does not exceed one-half ofthe desired step response time (SRT). Assuming that the rise time equalsabout 2.2 τ (as is typical for first order systems), and that the clockperiod is one-half the desired response time, then the cutoff frequencyof the filter 144 is selected to be on the order of 0.35 f_(CLK). Moreparticularly, T_(CLK)=½*SRT=½*(2*rise time)=2.2τ. Sincef_(cutoff)=1/(2*π*τ), we get 1/(2*π*(T_(CLK)/2.2) or f_(cutoff)=0.35f_(CLK). With such a cutoff frequency, the filter 144 will not totallyattenuate the ripple since the ripple contains harmonics of f_(CLK).However, the selective filter 150 does completely eliminate the rippleas will become apparent.

The selective filter 150 is a discrete time filter having zeros locatedat N(f_(SF)/2), where N is any integer and f_(SF) is the samplingfrequency. In one illustrative embodiment shown and described below inconjunction with FIG. 7, the selective filter 150 is a time domainaveraging filter with which the input signal 148 is averaged at the ratef_(CLK) and the sampling frequency f_(SF) is selected to be equal to twotimes the clock frequency f_(CLK). The resulting sensor output signal118 is shown in FIG. 4E to include only the magnetic signal component(identical to magnetic signal component 148 a as provided at the outputof the filter 144), since the selective filter entirely eliminates theoffset signal component 148 with its associated ripple.

As is apparent from consideration of FIG. 4, this embodiment contains aneven number of modulation circuits between the output of the Hall platemodulation circuit 50 that modulates the Hall offset signal componentand the input to the filter stage 120. Specifically, the modulationcircuit output signal 114 is processed by two modulation circuits 130and 140, before reaching the low pass filter 144. Modulation circuit 130demodulates the offset signal component to baseband and modulationcircuit 140 then up-converts the offset signal component so that theselective filter 150 can remove the offset and its associated ripple tothereby recover the desired magnetic signal component.

Referring to FIG. 5, an alternative Hall effect sensor 200 according tothe present invention includes a switched Hall plate 70 of the typeshown in FIG. 3 that provides a switched Hall plate output signal 214comprising a substantially invariant Hall offset signal component V_(op)and a modulated magnetic signal component V_(H), as shown in FIG. 5A.Also shown in dotted lines in FIG. 5A is the substantially invariantHall offset signal component V_(op). The sensor 200 further includes achopped amplifier 210 having an input coupled to the output of theswitched Hall plate 70 and an output at which an amplified signal 216 isprovided. A filter stage 220 has an input coupled to the output of theamplifier stage 210 and an output at which the sensor output signal 218is provided. Here again, the optional smoothing filter is not shown.

As in the embodiment of FIG. 4, the amplifier 210 is a closed loopamplifier having a feedback network 224, as shown. The illustrativefeedback network shown in FIG. 6 is suitable to provide the feedbacknetwork 224 in the embodiment of FIG. 5. It will be appreciated by thoseof ordinary skill in the art, that the same advantages described abovein connection with the closed loop amplifier 110 of FIG. 4 are realizedin the sensor of FIG. 5 (e.g., high linearity, gain stability, highergain and area savings).

As with the embodiment of FIG. 4, since the magnetic signal component atthe input to gain stage 238 is modulated, the tighter bandwidth requiredto implement the amplifier 210 in a closed loop form must be balancedwith the minimum bandwidth necessary to pass the desired magnetic signalcomponent. Since the modulated magnetic signal component does not gothrough the filter 244 or the Miller feedback stage 224, the closed loopbandwidth does not affect the modulated signal bandwidth. Only theamplifier section through which the modulated magnetic signal componentpasses needs to have enough bandwidth to pass the desired magneticsignal component.

A summing node 226 has inputs coupled to the feedback network 224 and anoutput coupled to a first modulation circuit, shown here in the form ofa pair of cross-coupled switches 230 that modulate the incoming signalat the clock frequency f_(CLK). The output signal of modulation circuit230 is coupled to an input of the gain stage 238. The switched Halloutput signal 214 is also coupled to the input of the gain stage 238, asshown. Thus, in this embodiment, the switched Hall output signal 214 isnot processed by the modulation circuit 230. Also shown at the input tothe gain stage 238 is a voltage source 234 representing the amplifieroffset signal component V_(oa).

Gain stage 238 provides an amplified signal to a further modulationcircuit, here in the form of a pair of cross-coupled switches 240, asshown. As noted, the gain stage 238 must have a bandwidth large enoughto pass the magnetic signal component that has been modulated by theswitched Hall plate 70. In one illustrative embodiment, the gain stagebandwidth is at least five times the clock frequency f_(CLK).

Modulation circuit 240 operates at the clock frequency f_(CLK) toprovide the amplified signal 216 having a demodulated magnetic signalcomponent and a modulated offset signal component, as shown in FIG. 5B.Also shown in dotted lines in FIG. 5B is the demodulated magnetic signalcomponent 216 a. Note again that the offset being modulated by thesecond modulation circuit 240 includes the Hall offset signal componentV_(op) (as represented by voltage source 78 in FIG. 3) and the amplifieroffset signal component V_(oa) (as represented by voltage source 234 inFIG. 5), which offsets are added at the input to the gain stage 238.

For reasons described above in connection with FIG. 4, the cutofffrequency of the filter 244 is selected to be on the order of 0.35f_(CLK). With such a cutoff frequency, the filter 244 will not totallyattenuate the ripple caused by the offset signal component since theripple contains harmonics of f_(CLK). However, the selective filter 250does completely eliminate the ripple. The output signal 248 of the lowpass filter 244 is shown in FIG. 5C. Also shown in dotted lines in FIG.5C, is the demodulated magnetic signal component 248 a of the signal248.

Here, the sampling frequency f_(SF) of the selective filter 250 isselected to be equal to two times the clock frequency f_(CLK), resultingin the filter zeros being located at f_(CLK) and its harmonics. Thus,since the selective filter 250 removes frequency components at f_(CLK)and its harmonics, the undesired output ripple, that has an amplitudeproportional to the DC offset signal component, and thus the offsetsignal component itself is eliminated. The resulting sensor outputsignal 218 is shown in FIG. 5D to include only the magnetic signalcomponent (identical to magnetic signal component 248 a as provided atthe output of the filter 244), since the selective filter entirelyeliminates the offset signal component.

As in the embodiment of FIG. 4, the selective filter 250 is a timedomain averaging filter with which the input signal 248 is averaged atthe rate f_(CLK). An illustrative embodiment for the selective filter250 is described and shown in connection with FIG. 7.

As is apparent from consideration of FIG. 5, this embodiment contains anodd number of modulation circuits between the output of the Hall platemodulation circuit 70 that modulates the magnetic signal component andthe input to the filter stage 220. Specifically, the modulation circuitoutput signal 214 is processed by one modulation circuit 240, beforereaching the low pass filter 244. Modulation circuit 240 up-converts theoffset signal component so that the selective filter 250 can remove theoffset and its associated ripple to thereby recover the desired magneticsignal component.

The modulation circuit 230 is shown in dotted lines to illustrate thatits position may be varied. More particularly, the modulation circuit230 may be positioned as shown in FIG. 5, between the summing node 226and the gain stage 238. Alternatively, the modulation circuit 230 may beprovided as part of the feedback network 224. In either position, themodulation circuit 230 modulates the feedback signal prior to its beingadded to the modulated magnetic signal component at the input to thegain stage 238.

Referring to FIG. 6, a portion of a Hall effect sensor 300 similar toHall effect sensor 200 of FIG. 4 is shown to include an illustrativefeedback network 310 of the type suitable to provide the feedbacknetwork 124 of FIG. 4 or the feedback network 224 of FIG. 5. The sensorportion 300 includes an amplifier 312, similar to amplifier 210 of FIG.4 and thus, including a summing node 314, a first modulation circuit,here shown in the form of a pair of cross-coupled switches 318, a gainstage 320, and a second modulation circuit, here shown in the form of apair of cross-coupled switches 324, all arranged and operable asdescribed in connection with similar respective elements 126, 130, 138,and 140 of FIG. 4. The sensor portion 300 further includes ananti-aliasing low pass filter 328 arranged and operable as described inconnection with similar filter 144 of FIG. 4.

The feedback network 310, like the feedback network 124 of FIG. 4, hasan input coupled to the output of the filter 328 and provides a feedbacksignal to an input of summing node 314. In some embodiments, it isdesirable to provide the feedback network 310 with the capability ofadjusting the gain of the amplifier 312. This feature is particularlyadvantageous in applications which provide sensitivity/gain trimming,for example in applications where airgaps need to be calibrated andadjusted to a certain range. To this end, preferably, the feedbacknetwork 310 includes an active amplifier, such as a transconductanceamplifier whose output current can be adjusted to a desired level. Asone example, the feedback network 310 may include a Gilbert cell whichis a current multiplier that can be used to adjust the gain of thefeedback amplifier 334 and thus, the overall closed loop gain. It willbe appreciated by those of ordinary skill in the art that other feedbacknetworks can be used to provide gain adjustment capabilities for thefeedback amplifier, such as by changing a resistor, a voltage or acurrent, thereby allowing the overall gain of the Hall effect sensor tobe adjusted or programmed.

More particularly, the feedback network 310 includes a first modulationcircuit, shown here in the form of a pair of cross-coupled switches 330having an input coupled to the output of filter 328 and an outputcoupled to a feedback gain stage 334. The output of the gain stage 334is coupled to an input of a further modulation circuit, also shown herein the form of a pair of cross-coupled switches 338, which switchesprovide at an output the feedback signal for coupling to the summingnode 314, as shown. The modulation circuits 330, 338 modulate therespective input signal at the clock frequency f_(CLK).

Given the active elements of the Gilbert cell network 310, it may bedesirable to eliminate the offset contribution from this network. Tothis end, the feedback amplifier 334 may be chopped, as provided in theembodiment of FIG. 6 with modulation circuits 330 and 338. With thisarrangement, the modulated offset coming from the feedback amplifier 334goes around the loop and also through the filter 328, so that thisoffset is treated in the same way as both the offset signal componentfrom the Hall plate and the offset signal component from the forwardamplifier 320.

It will be appreciated by those of ordinary skill in the art that thefeedback network 310 of FIG. 6 is one of various ways to implement thefeedback network. As one example, a resistive feedback network could beused.

Referring to FIG. 7, an illustrative selective filter 400 of the typesuitable for use as the selective filter 40 of FIG. 1, the selectivefilter 150 of FIG. 4, or the selective filter 250 of FIG. 5 includes afirst sample and hold circuit 404 and a second sample and hold circuit408, each having an input responsive to an anti-alias filtered signal,like signal 148 of FIG. 4 or signal 248 of FIG. 5. Each sample and holdcircuit 404, 408 has an output coupled to an averaging network 410, asshown. The output signal 412 of the averaging network 400 provides theoutput of the selective filter 400.

Referring also to FIG. 7A, in the frequency domain, the transferfunction 420 of the filter 400 is shaped like a sinc function, while inthe time domain is shaped like a rectangular function. The filter 400removes frequency components at N(f_(SF)/2), where N is any integer andf_(SF) is the sampling frequency and attenuates higher frequencycomponents. In the illustrative embodiment in which the samplingfrequency f_(SF) is equal to twice the clock frequency f_(CLK), thefilter removes components at f_(CLK) and its harmonics.

It is important to note that the selective filter 400 provides anegligible attenuation from DC up to approximately f_(SF)/8, with theattenuation increasing gradually from that frequency up to f_(SF)/2.Ideally, at a frequency of f_(SF)/2, the attenuation is infinite (i.e.,zero in the frequency domain), as shown in FIG. 7A. However, this doesnot pose a constraint on the system, assuming that the bandwidth for themagnetic signal component being processed is much smaller than f_(SF).While an input step signal, which might represent a fault condition in,for example, a battery current sensor application, would not have such abandwidth much smaller than f_(SF), this too is not problematic since,in this scenario, the system is not focused on providing an exactreplica of the input waveform but rather is focused on reacting fastenough in order to eventually trigger a comparator controlling thebattery supply connections.

Referring also to FIG. 7B signals 148 and 148 a of FIG. 4D are shownwith notations to illustrate operation of the selective filter 400. Inoperation, the first sample and hold circuit 404 samples the inputsignal in response to a sample clock signal 406 having a frequency equalto the modulation frequency f_(CLK) and a phase Φ, whereas the secondsample and hold circuit 408 samples the input signal in response to asample clock signal 412 at the same frequency f_(CLK), but at a phaseΦ+θ, where θ is equal to 180°. Described another way, the first sampleand hold circuit 404 samples the input signal at timest=t0+N*T_(SF)=t0+(N/2)*T_(CLK); whereas the second sample and holdcircuit 408 samples the input signal at timest=t0+(N+1)*T_(SF)=t0+((N+1)/2)*T_(CLK), where t0 is an arbitrary timeand N is an integer. Short pulses are used to perform the samplingoperation. The duration of the pulses is set large enough to allow thesignal to reach its final value before holding, as is a function of theRC time constant associated with the resistance of the sample and holdswitches and the capacitance of the capacitor. In one illustrativeembodiment, the pulse width is on the order of 200 ns.

For simplicity of illustration, the sample and hold circuit 404 is shownin FIG. 7B to sample the input signal at the peaks of the signal 148 andthe sample and hold circuit 408 samples the input signal at the valleysof the signal. Since the peaks of the input signal 148 correspond totransitions of the modulation clock signal, in the illustrated example,the sample clock signals coincide with transitions of the modulationclock signal. However, it will be appreciated that the phase shiftbetween the sample clock signals and the modulation clock signal can bearbitrary. In fact, it may be desirable to sample the input signalcloser to the zero crossing of the ripple in order to avoid large signalexcursions.

This synchronized sampling of the input signal at twice the ripplefrequency allows averaging the ripple signal, thereby completelyremoving it. Furthermore, the ripple average value is obtained afterjust one modulation clock period 1/f_(CLK) and thus, this is the onlydelay the selective filter introduces.

According to the selective filter operation described thus far, in whichthe input signal is averaged at the modulation frequency rate off_(CLK), several samples are accumulated, averaged, and then discardedin order to accumulate and average new samples to provide the nextaveraged signal value. This type of filter operation may be referred toas an “accumulation and dump” operation and may be described in thecontext of the illustrative circuit as the selective filter outputsignal comprising a plurality of signal averages, with each signalaverage being based on samples of the anti-aliasing filter output signaltaken within a single modulation clock cycle.

It will be appreciated by those of ordinary skill in the art howeverthat, alternatively, a running average may be used in which N samplesare stored and averaged to provide a first averaged signal value andwhen a new sample is taken (i.e., sample N+1), the oldest previouslystored sample (i.e., sample 1) is dropped and a new averaging isperformed based on the previously stored samples (i.e., samples 2, 3, .. . N) and the new sample (i.e., sample N+1). In this case, the inputsignal is averaged at the sampling rate f_(SF). In the context of theillustrative circuit, this type of running average operation may bedescribed as the selective filter output signal comprising a pluralityof signal averages, with each signal average being based on a pluralityof samples of the anti-aliasing filter output signal used to provide aprevious signal average and a new sample of the anti-aliasing filteroutput signal.

Advantageously, the selective filter 400 has the property of trackingany change on the clock frequency, for example as may be due totemperature or process variations. This is because the samplingfrequency f_(SF) is selected to be twice the clock signal frequencyf_(CLK) and is in fact generated from the clock signal. With thisarrangement, precise synchronization of the filter 400 to the ripplefrequency f_(CLK) is achieved, thereby ensuring accurate cancellation ofthe offset ripple.

As noted above in connection with FIG. 1, in addition to removing allthe signal components at the clock frequency f_(CLK) and its harmonics,the filter also attenuates the other frequency components above a givenbandwidth. The extent of attenuation at frequencies other than harmonicsof the clock frequency is a function of the number of samples of theinput signal taken within one clock period, with greater attenuation inthe filter sidelobes being achieved by taking more samples. Theparticular selection of the sample frequency requires a tradeoff betweengreater signal attenuation as is achieved with by taking more samples(i.e., a higher overall sampling frequency) versus area efficiencyachieved by taking fewer samples (i.e., a lower overall samplingfrequency). It will be appreciated by those of ordinary skill in the artthat the illustrative embodiment in which the sample frequency f_(SF) istwice the clock frequency f_(CLK) represents the minimum samplefrequency possible to achieve the selective filter averaging operation.

More generally, the selective filter may be designed to take N pairs ofsamples of the input signal during each clock cycle, again where 1 isthe minimum value of N. Samples are taken in pairs (i.e., an even numberof samples are taken during each clock cycle) in order to average outthe ripple during each clock cycle, which ripple is symmetrical aroundthe magnetic signal component. With this arrangement, the samplingfrequency f_(SF) is a multiple of the modulation frequency f_(CLK). Forproper selective filter operation, the clock signals controlling a givenpair of samples are separated in phase by 180 degrees and the clocksignals controlling different pairs of samples are arbitrarily separatedin phase.

Thus, the more general filter may be described as including a pluralityof sample and hold circuits arranged in pairs, each having an inputcoupled to the output of the anti-aliasing filter and an output.Specifically, the filter includes N pairs of sample and hold circuits,or 2N sample and hold circuits. The filter further includes an averagingcircuit having a plurality of inputs, each coupled to the output of arespective sample and hold circuit, and an output at which the selectivefilter output signal is provided. Each of the 2N sample and holdcircuits samples the low-pass filtered signal at the modulationfrequency f_(CLK) (so that the signal is sampled during each clock cycleat a multiple of the modulation frequency) and at a phase separated fromthe phase of the other sample and hold circuit of the same pair by 180degrees and a phase arbitrarily separated from the phase of the othersample and hold circuit pairs.

Referring to FIG. 8, a further alternative chopped Hall effect sensor500 according to the invention includes switched Hall plate 50 of thetype described in FIG. 3 that modulates the offset signal component. Thesensor 500 further includes a chopped amplifier 510 having an inputcoupled to the output of the switched Hall plate 50 and an output atwhich an amplified signal 516 is provided. A filter stage 520 has aninput coupled to the output of the amplifier stage 510 and an output atwhich the sensor output signal 518 is provided. Again, the optionalsmoothing filter is not shown.

The amplifier 510 is substantially identical to the amplifier 110 ofFIG. 4 and thus includes a summing node 526, a first modulation circuit,shown here in the form of a pair of cross-coupled switches 530, a gainstage 538, and a second modulation circuit also shown here in the formof a pair of cross-coupled switches 540, all arranged and operable inthe same manner as like respective components 126, 130, 138, and 140 inFIG. 4. The filter stage 520 is substantially identical to the filterstage 120 of FIG. 4 and thus, includes an anti-aliasing filter 544 and aselective filter 550 (which may be of the type shown in FIG. 7),arranged and operable in the same manner as like respective components144 and 150 in FIG. 4.

As in the embodiment of FIG. 4, the amplifier 510 is a closed loopamplifier having a feedback network 524, as shown. The illustrativefeedback network 310 shown in FIG. 6 is suitable for use to provide thefeedback network 524 in the embodiment of FIG. 8.

The sensor 500 of FIG. 8 differs from the sensor 100 of FIG. 4 in that,in FIG. 8, the amplifier loop is closed around the selective filter 550.Thus, the feedback network 524 has an input coupled to the output of theselective filter 550 and an output coupled to the summing node 526, asshown.

This arrangement is advantageous in embodiments in which the polesassociated with the amplifier loop are located so as to prevent thezeros introduced by the selective filter 550 from adversely impactingthe loop stability. In other words, with the selective filter 550positioned within the amplifier loop as shown in FIG. 8, the zerosintroduced by the selective filter (as will be located at harmonics ofthe sample frequency) will contribute to the loop stability. Again,depending on the location of the poles, this might not be problematic.And positioning the selective filter 550 within the amplifier loopprovides the benefit of minimizing the impact of the any offsetassociated with the selective filter on the overall system offset. Thisis because, with the selective filter positioned as shown in FIG. 8, itsoffset contribution is divided by the open loop gain, which is typicallyon the order of 60 to 100 dB, thereby making its contribution to theoverall system offset performance negligible. By contrast, in theembodiments of FIGS. 4 and 5, in which the respective selective filteris located outside of the amplifier loop, any offset associated with theselective filter may impact the overall offset performance,particularly, if the chopped amplifier gain is not large by comparison.

Referring to FIG. 9, a waveform 900 representing an illustrativemagnetic step disturbance is shown. FIG. 9A shows an illustrative stepresponse of the anti-aliasing filter of the inventive Hall effectsensor, for example of filter 38 of FIG. 1, filter 144 of FIG. 4, filter244 of FIG. 5 and filter 544 of FIG. 8, for a sensor having an amplifierbandwidth on the order of 250 KHz in response to the input stepdisturbance of FIG. 9. In particular, the output of the anti-aliasingfilter is shown in FIG. 9A both for the case where a Hall offset voltageexists (as labeled 904) and for the case where there is no Hall offsetvoltage (as labeled 908). And FIG. 9B shows an illustrative stepresponse 910 of the selective filter of the inventive chopped Halleffect sensor, for example of filter 40 of FIG. 1, filter 150 of FIG. 4,filter 250 of FIG. 5, and filter 550 of FIG. 8 in response to the inputstep disturbance of FIG. 9. As can be seen in FIG. 9, the selectivefilter introduces a delay of only one clock cycle.

In summary, the above-described Hall effect sensors 10 of FIG. 1, 100 ofFIG. 4, 200 of FIG. 5, and 500 of FIG. 8 provide an effective circuittopology to remove undesirable Hall and amplifier offset signalcomponents and provide the recovered magnetic signal component withoutripple, without fold-back noise and with a fast response time. Theselective filter averages the ripple, thereby completely removing it,without introducing significant delay (just 1/f_(CLK)) The anti-aliasinglow pass filter prevents noise fold-back, and a fast step response timeis achieved by proper design considerations including selection of theclock signal frequency f_(CLK) and the anti-aliasing filter cutofffrequency.

All references cited herein are hereby incorporated herein by referencein their entirety.

Having described preferred embodiments of the invention, it will nowbecome apparent to one of ordinary skill in the art that otherembodiments incorporating their concepts may be used.

For example, while the Hall effect sensor embodiments described hereinhave the Hall plate modulation circuit operating at the same modulationfrequency as the signal modulation performed by the amplifier stage, itwill be appreciated by those of ordinary skill in the art that, incertain instances, it may be desirable to modulate the Hall offsetsignal component at one frequency and the amplifier offset at adifferent frequency. In this case, the modulation frequencies must befar enough separated and the selective filter must be tuned to bothfrequencies in order to properly demodulate the signals and keep themseparated.

It will also be appreciated by those of ordinary skill in the art thatHall effect sensors according to the invention may include more tan one(i.e., N) Hall plates for providing respective current or voltage outputsignals in various arithmetic combinations of the sensed magnetic field.In this case, N modulation circuits are provided, each processing theoutput signal of a respective Hall plate and providing an output signalfor coupling to a summing node for further processing as described abovein connection with the various embodiments of the invention.

It is felt therefore that these embodiments should not be limited todisclosed embodiments, but rather should be limited only by the spiritand scope of the appended claims.

1. A Hall effect sensor comprising: a Hall element having an output atwhich is provided a Hall output signal that varies in response to amagnetic field, the Hall output signal comprising a magnetic signalcomponent and an offset signal component; a Hall plate modulationcircuit having an input responsive to the Hall output signal and havingan output at which is provided a modulation circuit output signal,wherein the Hall plate modulation circuit is operable to modulate themagnetic signal component or the offset signal component at a modulationfrequency f_(CLK); an amplifier having an input responsive to the Hallplate modulation circuit output signal and an output at which isprovided an amplifier output signal; and a filter having an inputresponsive to the amplifier output signal and an output at which isprovided a sensor output signal, wherein the filter comprises ananti-aliasing filter and a discrete time selective filter tuned at themodulation frequency to remove the offset signal component, wherein theanti-aliasing filter provides a bandwidth limited signal for filteringby the selective filter, wherein the anti-aliasing filter removesfrequency components above a predetermined frequency selected to preventaliasing by the discrete time selective filter.
 2. The Hall effectsensor of claim 1 wherein the amplifier is a chopped amplifier.
 3. TheHall effect sensor of claim 2 wherein the amplifier is chopped at themodulation frequency.
 4. The Hall effect sensor of claim 2 wherein theamplifier is chopped at a different frequency than the modulationfrequency.
 5. The Hall effect sensor of claim 1 wherein the amplifier isa closed loop amplifier.
 6. The Hall effect sensor of claim 5 whereinthe amplifier comprises a feedback network that is adjustable to adjustthe gain of the amplifier.
 7. The Hall effect sensor of claim 1 whereinthe Hall plate modulation circuit is operative to modulate the offsetsignal component and wherein the Hall effect sensor comprises an evennumber of modulation circuits between the output of the Hall platemodulation circuit and the input of the filter.
 8. The Hall effectsensor of claim 7 wherein the amplifier comprises: a summing node havinga first input coupled to the output of the Hall plate modulationcircuit, a second, feedback input, and an output; a first modulationcircuit having an input coupled to the output of the summing node and anoutput; a gain stage having an input coupled to the output of the firstmodulation circuit and an output; a second modulation circuit having aninput coupled to the output of the gain stage and an output; and afeedback network having an input coupled to the filter and an outputcoupled to the second input of the summing node.
 9. The Hall effectsensor of claim 8 wherein the input of the feedback network is coupledto a selected one of an input of the anti-aliasing filter or an outputof the anti-aliasing filter.
 10. The Hall effect sensor of claim 8wherein the input of the feedback network is coupled to an output of theselective filter.
 11. The Hall effect sensor of claim 8 wherein thefeedback network is chopped.
 12. The Hall effect sensor of claim 1wherein the Hall plate modulation circuit is operative to modulate themagnetic signal component and wherein the Hall effect sensor comprisesan odd number of modulation circuits between the output of the Hallplate modulation circuit and the input of the filter.
 13. The Halleffect sensor of claim 12 wherein the amplifier comprises: a summingnode having an input and an output; a first modulation circuit having aninput coupled to the output of the summing node and an output; a gainstage having an input coupled to the output of the Hall plate modulationcircuit and to the output of the first modulation circuit and an output;a second modulation circuit having an input coupled to the output of thegain stage and an output coupled to the filter; and a feedback networkhaving an input coupled to the filter and an output coupled to the inputof the summing node.
 14. The Hall effect sensor of claim 13 wherein theinput of the feedback network is coupled to a selected one of an inputof the anti-aliasing filter or an output of the anti-aliasing filter.15. The Hall effect sensor of claim 13 wherein the input of the feedbacknetwork is coupled to an output of the selective filter.
 16. The Halleffect sensor of claim 12 wherein the amplifier comprises: a summingnode having an input and an output; a gain stage having an input coupledto the output of the Hall plate modulation circuit and to the output ofthe summing node and an output; a first modulation circuit having aninput coupled to the output of the gain stage and an output coupled tothe filter; and a feedback network having an input coupled to the filterand an output coupled to the input of the summing node, wherein thefeedback network is chopped.
 17. The Hall effect sensor of claim 16wherein the input of the feedback network is coupled to an input of theanti-aliasing filter.
 18. The Hall effect sensor of claim 16 wherein theinput of the feedback network is coupled to an output of theanti-aliasing filter.
 19. The Hall effect sensor of claim 16 wherein theinput of the feedback network is coupled to an output of the selectivefilter.
 20. The Hall effect sensor of claim 1 wherein the anti-aliasingfilter has an input responsive to the amplifier output signal and anoutput at which a low-pass filtered signal is provided and wherein theselective filter has an input coupled to the output of the anti-aliasingfilter and an output at which a selective filter output signal isprovided and wherein the selective filter comprises: a plurality ofsample and hold circuits arranged in pairs, each sample and hold circuithaving an input coupled to the output of the anti-aliasing filter and anoutput; and an averaging circuit having a plurality of inputs eachcoupled to the output of a respective one of the sample and holdcircuits and an output at which the selective filter output signal isprovided, wherein each of the sample and hold circuits samples thelow-pass filtered signal at the modulation frequency and at a phaseseparated from the phase of the other sample and hold circuit of thesame pair by 180 degrees and at a phase arbitrarily separated from thephase of the sample and hold circuits of the other pairs.
 21. The Halleffect sensor of claim 20 wherein the selective filter output signalcomprises a plurality of signal averages, each signal average beingbased on samples of the anti-aliasing filter output signal taken withina single cycle of a modulation clock signal having the modulationfrequency.
 22. The Hall effect sensor of claim 20 wherein the selectivefilter output signal comprises a plurality of signal averages, eachsignal average being based on a plurality of samples of theanti-aliasing filter output signal used to provide a previous signalaverage and a new sample of the anti-aliasing filter output signal. 23.The Hall effect sensor of claim 1 wherein the anti-aliasing filter hasan input responsive to the amplifier output signal and an output atwhich a low-pass filtered signal is provided and wherein the selectivefilter has an input coupled to the output of the anti-aliasing filterand an output at which a selective filter output signal is provided andwherein the selective filter comprises: a first sample and hold circuithaving an input coupled to the output of the anti-aliasing filter and anoutput; a second sample and hold circuit having an input coupled to theoutput of the anti-aliasing filter and an output; and an averagingcircuit having inputs coupled to the outputs of the first and secondsample and hold circuits and an output at which the selective filteroutput signal is provided, wherein the first sample and hold circuitsamples the low-pass filtered signal at times t=t0+N·T_(SF) and whereinthe second sample and hold circuit samples the low-pass filtered signalat times t=t0+(N+1)·T_(SF), wherein t0 is an arbitrary time, N is aninteger and T_(SF) is 1/(2·f_(CLK)), where f_(CLK) is the modulationfrequency.
 24. The Hall effect sensor of claim 1 wherein the filterfurther comprises a smoothing filter.
 25. The Hall effect sensor ofclaim 1 further comprising: a plurality of Hall elements, each having anoutput at which is provided a Hall output signal that varies in responseto a magnetic field; a plurality of modulation circuits, each having aninput responsive to the Hall output signal from a respective one of theplurality of Hall elements and an output at which is provided amodulation circuit output signal; and an element responsive to each ofthe modulation circuit output signals for providing the modulationcircuit output signal to the amplifier as a mathematical combination ofthe plurality of modulation output signals.